Philips Stereo Amplifier AN1651 User Manual

INTEGRATED CIRCUITS  
AN1651  
Using the NE/SA5234 amplifier  
Author: Les Hadley  
1991 Oct  
Philips  
Semiconductors  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
V
V
IN  
IN  
V
CC  
V
V
V
OUT  
CC  
V
CC  
V
OUT  
47k  
47k  
5V  
5V  
+
V
IN  
V
t
GND  
GND  
t
V
OUT  
V
GND  
CONVENTIONAL OP AMP  
PHILIPS NE5234  
SL00569  
Figure 2. Output Inversion Protection  
non-inverting input. The output is taken from multiple collectors on  
the non-inverting side and provides matching for the following stage.  
6
5
4
3
2
Class-AB control of the output stage is achieved by Q61 and Q62  
with the associated output current regulators. These act to monitor  
the smallest current of the non-load supporting output transistor to  
keep it in conduction. Thus, neither Q71 or Q81 is allowed to cutoff  
but is forced to remain in the proper Class-AB region.  
“N-MODE”  
CMRR  
“LARGE  
SIGNAL”  
CMRR  
V
+1 < V < V  
EE  
CM CC  
Overload protection is provided by monitor circuits consisting of  
R76-D2 for sinking and R86-D3 for sourcing condition at the output.  
When the output current, source or sink, reaches 15 milliamperes,  
drive current to the stage is shunted away from current sources IB6  
or IB9 reducing base current to driver transistors Q72 and Q82  
respectively.  
1
“N-MODE”  
< V +0.5V  
0.5  
V
< V  
CM  
CMRR  
EE  
EE  
V
EE  
V
OS  
mV  
-1  
NE5234 Common-Mode Operating Regions  
SL00631  
The prevention of saturation in the output stage is achieved by  
saturation detectors Q78 and Q88. When either Q71 or Q81  
approaches saturation, current is shunted away from the driver  
transistors, Q72 or Q83 respectively.  
Figure 3.  
For negative going input signals, which drive the inputs toward the  
rail and below, another set of diode-connected transistors come  
V
EE  
into operation. These steer the current from the input into Q8 or Q9  
emitter circuits again preventing the reversal effect.  
III. CHARACTERISTICS  
Figure 3 shows graphically how the N and P mode transitions relate  
to the common-mode input voltage and the offset voltage V  
.
OS  
Internal Frequency Compensation  
The use of nested Miller capacitors C2 through C6, in the  
intermediate and output sections, provides the overall frequency  
compensation for the amplifier. The dominant pole setting capacitor,  
C2, provides a constant 6dB/octave roll-off to below the unity gain  
frequency of 2.5MHz. Figure 5 shows the measured frequency  
response plot for various values of closed-loop gains.  
Intermediate Amplifier and Output Stage  
(Figure 4)  
The intermediate stage is isolated from the input amplifier by emitter  
followers  
to prevent any adverse loading effect. This stage adds gain to the  
over all amplifier and translates levels for the following class-AB  
current-control driver. Note that I is the inverting input and I the  
2
1
3
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
V
CC  
D2  
R82  
R85  
Q85  
R76  
Q83  
Q81  
I
I
B5  
B6  
I
B2  
I
B4  
I
B8  
Q82  
Q53,54  
Q51,52  
I
2
Q84  
C5  
C3  
INPUT  
+
OUTPUT  
C4  
C6  
D3  
Q72  
C2  
B4  
I
1
Q78  
Q71  
Q61  
Q62  
Q75  
V
CLASS  
AB  
CONTROL  
C1  
I
B3  
R86  
R75  
I
B9  
I
B7  
V
EE  
INTERMEDIATE STAGE  
CURRENT CONTROL  
CLASS AB OUTPUT  
SL00632  
Figure 4.  
100  
80  
60  
dB  
G1000  
40  
20  
0
10Hz  
100Hz  
1kHz  
10kHz  
100kHz  
1MHz  
6
10 @ 10  
FREQUENCY  
SL00633  
Figure 5. NE5234 Closed Loop Gain vs Frequency  
4
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
typically 0.2pA/Hz. The 1/f region was not determined for either  
current or voltage noise.  
+2.5V  
4
–2.5V  
E
for R = 10-nV/Hz  
n
S
11  
22  
HP  
+
5234  
3585  
nV  
SPECTRUM  
ANALYZER  
Hz  
19  
95%  
INT.  
600Ω  
100  
10Ω  
47k  
x10  
18  
17  
16  
SL00634  
Figure 6. Noise Test Circuit  
100 200  
2000  
10000  
a.  
IV. NOISE REFERRED TO THE INPUT  
pA Hz  
The typical spectral voltage noise referred to each of the op amps in  
the NE/SA5234 is specified to be 25nV/Hz. Current noise is not  
specified. In the interest of providing a balance of information on the  
device parameters, a small sample of the standard NE5234s, were  
tested for input noise current. While this data does not represent a  
specification, it will give the designer a ball park figure to work with  
when beginning a particular design with the device. For  
completeness I have provided the corresponding spectral noise  
voltage data for the same sample. The data was taken using an  
HP3585A spectrum analyzer which has the capability of reading  
noise in nV/Hz.  
0.5  
12  
in 10  
P
0.1  
100  
f
1k  
10000Hz  
200  
2k  
P
b.  
SL00635  
Figure 7. Typical Noise Current and Voltage vs Frequency  
The test circuit is shown in Figure 6. As is typical for such  
measurements the amplifier under test is terminated at its input first  
with a very low resistance, for the voltage noise reading, followed by  
the same test with a high value of resistance to register the effect of  
current noise. The amplifier is set to a non-inverting  
V. GUIDE LINES FOR MINIMIZING NOISE  
When designing a circuit where noise must be kept to a minimum,  
the source resistances should be kept low to limit thermally  
generated degradation in the overall output response.  
closed-loop gain of 20dB. Dual supply operation was chosen to  
allow direct termination of the input resistors to ground.  
Orders-of-magnitude should be kept in mind when evaluating noise  
performance of a particular circuit or in planning a new design. For  
instance, a transducer with a 10ksource resistance will generate  
2µV of RMS noise over a 20kHz bandwidth. Using the graphical  
data above, total noise from a gain stage may be calculated.-  
The measurements were made over the range from 200Hz to 2kHz.  
Each sample is measured at 200Hz, 500Hz, 1kHz and 2kHz. The  
data is averaged for each frequency and then the small sample  
distribution is derived statistically giving the standard deviation  
relative to the mean.  
(EQ. 1.)  
Amplifier Noise Voltage  
Referring to the graph in Figure 7a, the equivalent voltage noise is  
seen to average 18 nV/Hz. The 95% confidence interval is  
determined to be approximately one nV/Hz. The majority of the  
errors which contribute to this measurement are due to the thermal  
noise of the parallel combination of the feedback resistor network, in  
addition to the 10termination resistor on the non-inverting input.  
At 300° Kelvin a 10resistor generates 0.4 nV/Hz and the  
feedback network’s equivalent resistance of 90generates  
1.2nV/Hz. Their order-of-magnitude difference from the main noise  
sources allows them to be neglected in the overall calculation of  
total stage noise.  
25nV Hz @ BW  
3.5mVRMS  
BW  
10kHz  
Noise from source 10kResistance –  
(EQ. 2.)  
(EQ. 3.)  
Noise Voltage from source resistance  
14nV Hz @ BW  
20mVRMS  
Current generated noise  
0.2pA Hz @ 103 @ BW  
0.28mVRMS  
Noise current is measured across a 47kresistor and averaged in  
the same manner. The thermal noise generated by this large  
resistance is not insignificant. At room temperature it is 28nV/Hz  
and must be subtracted from the total noise as measured at the  
output of the op amp in order to arrive at the equivalent current  
generated noise voltage. Figure 7b shows the derived current  
noise distribution for the small sample of 10 NE5234 devices. The  
result shows that noise current in the 200Hz to 2kHz frequency is  
The total noise is the root-to-sum-of-the-squares of the individual  
noise voltages –  
(EQ. 4.)  
En  
(3.5)2  
(2.0)2  
(0.28)2  
4.04mVRMS  
To determine the signal-to-noise ratio of the stage we must first  
choose a stage gain, make it 40dB, and a signal voltage magnitude  
from the transducer which we will set at 10mV  
. The resulting  
RMS  
5
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
signal-to-noise ratio at the output of this stage is determined by first  
3
(EQ. 7.)  
100x10  
1.6x10  
S N  
20 log10  
multiplying the gain times the signal which gives 1V  
with a  
4
RMS  
resultant noise of 400mV  
as  
. The signal-to-noise ratio is calculated  
RMS  
56dB  
4
A 56dB S/N will provide superior voice channel communications .  
(EQ. 5.)  
S N 20log10 (1.0 4x10  
)
68dB  
This is quite adequate for good quality audio applications.  
Next assume that the bandwidth is cut to 3.0kHz with an input of  
1mV . The RMS noise is modified by the ratio of the root of the  
1.6µV  
e
n
100kΩ  
e
n
RMS  
+
+
1mV  
noise channel bandwidths.  
RMS  
R
= 100Ω  
S
1mV  
SIGNAL  
RMS  
+1.6mV  
3x103  
20x103  
NOISE  
RMS  
x100  
(EQ. 6.)  
@ EN  
1.6mVRMS  
x10  
SL00636  
Amplified Noise = 160µV  
RMS  
Figure 8.  
+
V
CC  
10k  
1µF  
UNITY GAIN  
100k  
V
CC  
2
3
2
+
1
2.2µF  
600Ω  
R
= 600Ω  
L
1µF  
+
V
CC  
10k  
10k  
V
CC  
2
ST1700  
4.7µF  
3
2
+
1
1µF  
ST1700  
600Ω  
10k  
100pF  
1k  
DISTORTION  
ANALYZER  
40dB CIRCUIT  
SL00637  
Figure 9. NE5234 THD Test Circuits  
instance, a signal input which exceeds the input noise of the  
following stage by a factor of 10:1 will only be degraded by 0.5% or  
-46dB, neglecting the first-stage noise. If we use the preceding  
VI. MULTIPLE STAGE CONSIDERATIONS  
Since multiple noise generators are non-coherent, their total effect is  
the root-of-the-sum-of-the-squares of the various noise generators  
at a given amplifier input.  
example with a first-stage output signal of 100mV  
and a 56dB  
RMS  
S/N, and an output noise of 0.16mV. Following this with a 10kHz  
band limited gain-of-10 second-stage, with a 100knoise source at  
the non-inverting input, the combined S/N is calculated as follows:  
(assume a 100source resistance from amplifier #1)  
This makes orders-of-magnitude lower noise sources less important  
than the higher magnitude source. Therefore, when considering the  
combined signal-to-noise of multiple stages of gain, the first stage in  
a chain dominates making its design parameters the most critical.  
For this reason it is good practice to make the preamp stage gain as  
high as practical to boost signal levels to the second stage allowing  
at least an order-of-magnitude above the second-stage noise. For  
The Second stage output noise is:  
6
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
A series of tests are shown to allow you to see just how resistant  
this device is to generating clipping distortion. Two different gain  
configurations were chosen to demonstrate this particular feature:  
unity gain non-inverting and 40dB non-inverting. The test set-up  
was as shown in Figure 9. The Harmonic Distortion analyzer used  
to make the measurements was a Storage Technology ST1700.  
The test frequency is 1kHz. For single supply operation, as  
previously covered, the amplifier should be biased to half the supply  
voltage to minimize distortion. Operation with dual supplies is  
simpler from a parts count standpoint as isolation capacitors are not  
required. Also the time constants associated with charging and  
discharging these is eliminated . Figure 10a,b and c shows the total  
harmonic distortion in percent versus input voltage level at 1kHz in  
2
(EQ. 8.)  
(EQ. 9.)  
3 2  
(0.163x10  
)
4KT @ 100 @ 10, 000  
@ 10  
1.6mV  
K
Boltzman sConstant  
1.38x10  
Joule  
23  
DegKelvin  
T
300oK ; BW  
10kHz  
The amplified output signal = 1V  
RMS  
1
(EQ. 10.)  
S N  
20 log10  
56dB  
3
1.6x10  
V
RMS  
for a non-inverting, unity gain NE5234. The load on the  
amplifier output is 10k. Beginning with a supply voltage of 1.8V  
and an input level of 0.1V , distortion is well below 0.2% ad  
Note that there is no effect from the second-stage thermally  
generated resistor noise due to the dominating effect of the  
RMS  
remains there up to an input level just over 0.5V  
(1.4V ) and  
P-P  
RMS  
first-stage amplified noise being much greater than the input noise of  
the second-stage. In addition the equivalent noise resistance of the  
second-stage is essentially the output resistance of the first-stage  
plus any series resistance used in coupling the two. This is the  
parallel combination of source resistance with input terminating or  
biasing resistance.  
increases to 0.4% for for 0.6V  
(1.7V ).  
P-P  
RMS  
For a 2V supply, the input levels increase to 0.65V  
and  
RMS  
0.7V  
, respectively for similar levels of distortion. With a supply  
RMS  
voltage of 3.0V the input may be increased to 1V  
before THD  
RMS  
rises to 0.2% and 1.1V  
for only 0.8% THD. Operation with a  
RMS  
600load will only raise the THD figures slightly . By way of  
comparison, Figure 10c shows the greatly reduced dynamic range  
experienced when an LM324 is plugged into the test socket in place  
of the NE5234. Note that The THD is completely off scale for the  
case of 1.8 and 2.0V supply, then is barely usable for the low level  
end of the 3.0V supply example. Figure 11a, b, and c demonstrates  
the effect on harmonic distortion when closed loop gain is increased  
to 40dB in the non-inverting mode. It is evident that little increase in  
THD levels result. The graphs for the 2.0 and 3.0V supply case also  
include additional information on the effect of a 600load on  
distortion.  
VII. LOW HARMONIC DISTORTION  
The NE/SA5234 is extremely well adapted to reducing harmonic  
distortion as it relates to signal level and head room in audio and  
instrumentation circuits. Its unique internal design limits overdrive  
induced distortion to a level much below that experienced with other  
low voltage devices. As will be shown, the device is capable of  
operating over a wide supply range without causing the typical  
clipping distortion prevalent in companion operational amplifiers of  
this class.  
UNITY GAIN  
UNITY GAIN  
UNITY GAIN  
0.8  
3
3
LM324  
V
= 2.0V  
CC  
V
= 1.8V  
CC  
V
= 3.0V  
NE5234  
CC  
0
0
2
0
0.1  
1.1  
0.1  
1.0  
0.1  
1.1  
V
V
V
SL00638  
a.  
b.  
c.  
Figure 10. THD vs Supply Voltage for 1V  
Output  
RMS  
2.5  
3
V
= 2V  
CC  
= 10k/600Ω  
V
= 3.0V  
CC  
GAIN = 40dB  
THD for V  
= 1.8V  
CC  
-R = 10k/600Ω  
R
L
L
R
= 600Ω  
R
= 600Ω  
L
L
R
= 10kΩ  
L
P
R
= 10kΩ  
R
= 10kΩ  
L
L
0
0.1  
V
0.9  
0
0
0.1  
V
1.1  
0.1  
1.1  
a. V  
b.  
c.  
SL00639  
Figure 11. THD vs Load  
7
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
aforementioned factors that affect the signal-to-noise ratio of the  
VIII. GAIN-BANDWIDTH VS CLOSED LOOP FRE-  
stage and optimizing the Loop-gain. For example, a voice-band  
audio stage which requires 3kHz bandwidth, should be limited to a  
closed-loop gain of 40dB for lowest distortion in the output signal.  
For higher quality audio applications requiring a 20kHz bandwidth,  
the closed-loop gain must be limited to 20dB. This results in a  
Loop-gain of 20dB at the highest signal frequency.  
QUENCY RESPONSE  
Figure 5 shows the small signal frequency response of the NE5234  
versus closed-loop gain in dB. The test circuit is shown in Figure 6.  
The plot is taken from measured data and thus shows how each  
value of closed-loop gain coincides with the open-loop response  
curve. The NE/SA5234’s open-loop gain response has a uniform  
6dB/octave roll-off which continues beyond 2.5MHz. This factor  
guarantees each op amp in the IC a high stability in virtually any  
gain configuration. In making these measurements, dual supplies of  
±2.5V were used in order to allow a grounded reference plane and  
no coupling capacitors which might cause frequency related errors.  
A second consideration in the list of frequency dependent  
parameters is the effect of amplifier slew rate. Not only is it  
frequency dependent but it is also a function of signal amplitude, as  
we shall see in the next section.  
A
OL  
A critical parameter which affects the reproduction quality of  
complex waveforms is the gain-bandwidth-product of the operational  
amplifier. Essentially, this is a measure of the maximum frequency  
handling characteristics of any operational amplifier for a given  
closed-loop gain. As is evident from the graph, the NE/SA5234 has  
a 2.5MHz unity gain cross-over frequency — much higher than most  
other low voltage op amps. For comparison, the µA741 has a  
gain-bandwidth-product of 1MHz, as do the LM324 and the  
MC3403.  
-6dB/Octave  
LOOP  
GAIN  
A
CL  
f
f
u
S
SL00640  
Figure 12.  
IX. LOOP-GAIN  
The dynamic signal response of any closed-loop amplifier stage is a  
function of the Loop-gain of that particular stage. Loop-gain is equal  
to the open-loop gain in dB, at a given frequency, minus the  
closed-loop gain of the stage. The greater the Loop-gain, the lower  
the transfer function error of the device. Essentially, any parametric  
error is reduced by the factor of the Loop-gain. This includes output  
resistance and output signal voltage accuracy. It is good practice  
then to maximize Loop-gain to the degree that stage gain may be  
sacrificed for bandwidth. In some cases it is actually better to use  
two stages of gain in order to preserve signal quality than to use one  
high gain stage. Of course, there is a trade-off between the  
X. SLEW RATE RESPONSE  
The slew rate of an operational amplifier determines how fast it can  
respond to a signal, and is measured in volts-per-microsecond. The  
NE5234 has a typical slew rate of 0.8V/µs. Let us see just what this  
means in terms of signal handling capability. If a sinusoidal input  
signal, V , is used as reference, it is specified by its frequency and  
S
peak amplitude, V as follows:  
P
(EQ. 11.)  
VS  
VP sin (2 f t)  
2
V
V
= 1.096V  
= 630mV  
PK  
PK  
V
= 100mV  
PK  
0.02  
2000  
2000000  
(Hz)  
SL00641  
Figure 13. Slew Rate Limiting Amplitude vs Frequency  
Slew Rate (SR) is the time-rate-of-change of the signal voltage  
during any complete cycle, that is over the range of 0 to 2π. This  
amounts to taking the time derivative of the sine wave which results  
in multiplying the cosine by the factor ‘2πf’.  
Figure 13, the maximum allowable amplitude signal which can be  
reproduced is determined by the slew rate response line which gives  
peak output volts versus frequency in Hertz.  
Mathematically, slew rate is determined, by the equation below, as  
the derivative of the sine wave signal. The resultant slew rate  
function changes with both frequency and amplitude.  
An example of the trade off between signal amplitude and frequency  
is shown below for the NE5234 slew rate of 0.8V/µs. As shown in  
8
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
800,000 V/sec / 2π 1.414 volts peak = 90,090Hz. A graphical  
Slew Rate  
VP (2 f) cos (2 f t)  
representation of this relationship is shown in Figure 13. By using  
this graph along with the information in the preceding Figure 10 and  
Figure 11, which relate usable signal levels versus power supply  
voltage, the dynamic behavior of a particular design may be  
predicted. For instance, given a single supply configuration  
operating at 2.0V, Figure 10b shows an upper limit to input  
Note that maximum slew rate occurs where the input sine wave  
signal crosses the values of 0, π, and 2π on the radian axis. To get  
a feel for what this means in regards to the typical low voltage  
circuit, let us consider a 1V  
sinusoidal input to a unity gain  
RMS  
amplifier. The peak voltage in the above equation is 1.414V. One  
can then calculate the required slew rate to faithfully reproduce this  
signal for various signal frequencies. Or with a given slew rate and  
a required peak signal amplitude, the maximum frequency before  
slew rate limiting occurs may be determined. For example using the  
amplitude of 0.7V  
, or about 1V peak for 1% THD. Using this  
RMS  
level with the data in Figure 13 leads to a figure of 116kHz as an  
upper frequency limit for a unity gain amplifier stage operating at 2V  
DC.  
above amplitude of 1V  
, and the slew rate of the NE5234 which  
RMS  
is 800,000V/sec, one determines that the highest frequency  
component which may be reproduced before slew rate distortion  
occurs is:  
dVS  
(EQ. 12.)  
VP cos  
t
d t  
Slew Rate  
+
V
CC  
4
V
R
CC  
2
R
S
A
+
A
+
A
+
4
A
1
R
V
C
S
CC  
R
S
2
C
L
R
f
V
INPUT ISOLATION  
IN  
SL00642  
Figure 14. Single Supply Biasing in Cascade  
stage gain. Second stage biasing may now be provided by the  
output voltage of the first stage if non-inverting operation is used in  
the former. For lowest noise in a high gain input stage, the  
magnitude of the input source resistance is critical; low values of  
resistance are preferred over high values to minimize thermally  
generated noise.  
XI. PROCEDURES  
Single Supply Operation  
When the NE/SA5234 is used in an application where a single  
supply is necessary, input common-mode biasing to half the supply  
is recommended for best signal reproduction. Referring to Figure  
14, a simplified inverting amplifier input stage is shown with the  
simplest form of resistive divider biasing. The value of the divider  
resistance R is not critical and may be increased above the 10kΩ  
value shown as long as the bias current does not interfere with  
accuracy due to DC loading error. However the divider junction  
must be kept at a low AC impedance This is the purpose of bypass  
Non-Inverting Stage Biasing  
Non-inverting operation of an amplifier stage with single supply is  
similar to the previous example but the bias resistor R must now be  
S
sufficiently high to allow the signal to pass  
without significant attenuation. The input source resistance reflects  
the output resistance of the preceding stage or other sourcing  
device such as a bridge circuit of relatively high impedance. A  
simple rule of thumb is to make the bias resistor an order of  
magnitude larger than the generator resistance. Again the feed  
back network must be terminated capacitively. In this case R1 and  
capacitor C . Its use provides transient suppression for signals  
S
coming from the supply bus. A low cost 0.1µF ceramic disk or chip  
capacitor is recommended for suppressing fast transients in the  
microsecond and sub-microsecond region.  
Foil capacitors are simply too inductive for any high frequency  
bypass application and should be avoided. If low frequency noise  
such as 60Hz or 120Hz ripple is present on the supply bus, an  
electrolytic capacitor is added in parallel as shown. The  
common-mode input source resistance, R , should also be matched  
within a reasonable tolerance for maximizing the rejection of induced  
AC noise.  
the generator resistance should be matched and then R is matched  
S
to the feedback resistance ,R .  
F
In all cases proper bypassing of the NE5234 supply leads (Pins 4  
and 11) is very important particularly in a high noise environment.  
Bypass capacitors must be of ceramic construction with the shortest  
possible leads to keep inductance low. Chip capacitors are superior  
in this respect complimenting the increased use of surface mounted  
integrated devices. Note that both the NE5234D and the automotive  
grade SA5234D are available and are the surface mount versions of  
the device.  
S
The output of the first stage is now fixed at the common mode bias  
voltage and the amplified AC signal is referenced to this constant  
value. Capacitive coupling to the inverting input is of course  
required to prevent the bias voltage from being multiplied by the  
9
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
R
R
R
V
CC  
10k  
V
CC  
2
RECEIVE UNIT  
NE/SA5234  
4
A1  
+
T
S
R
A
N
S
+
A2  
CC  
4700Ω  
+
R
= 250Ω  
SH  
11  
F
R1  
C1  
R
SL00644  
SL00643  
Figure 15. Non-Inverting Biasing  
Figure 16. A 4-20mA Current Loop  
12k  
1.2M  
V2  
4.3k  
4.3k  
+5.0V  
O
|V – V |  
V
O
2
1
4
2
3
S.G.  
S.G.  
5.9mV  
25.6mV  
46.6mV  
0.5V  
2.50V  
4.63V  
+
+5.0V  
V
1
4.3k  
4.3k  
1k  
11  
S.G.: Matched Strain Gauge elements  
12k  
1.2MΩ  
SL00645  
Figure 17. Strain Gauge Amplifier  
+V  
CC  
1.2M  
4.3k  
4.3k  
4
S.G.  
S.G.  
2
+
SIGNAL  
COM  
12kΩ  
1
3
4.3k  
4.3k  
11  
Two-wire, Twisted-pair  
Shielded Line  
12kΩ  
1.2MΩ  
SL00646  
Figure 18. Remote Strain Gauge  
4-6V DC  
+
+3V  
+
CMOS  
+
4V  
VR  
SL00647  
Figure 19. Solar Regulator  
10  
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
drive the current loop. The sensitivity is actually in mA/V, or  
APPLICATIONS EXAMPLES  
Instrumentation  
transconductance, which is equal to 1/R . This sensitivity in this  
SH  
particular example is set to 4mA/V. Thus, with a bridge amplifier  
having a differential gain of 100, an input of 10mV will produce a  
4mA output current and 50mV will produce a 20mA output. Of  
course the line resistance plus receiver resistance must be within  
the voltage compliance range of the supply voltage to guarantee  
linear operation over the total range. A negative supply may be  
used if it is preferred to have the current loop referenced to ground.  
Strain Gauge Bridge Amplifier  
The circuit below shows a simple strain gauge circuit with a gain of  
100 (40dB) and operated from a single supply. The chart illustrates  
the transfer function of the circuit for a single order-of-magnitude  
signal differential range from the bridge beginning with 5mV up to  
50mV. The circuit is operated from a single 5V supply, but could  
equally as well be configured to use a dual balanced supply. It is  
immediately evident that the wide common-mode output range of  
the NE5234 is very advantageous in handling this wide range of  
signals with good linearity due to this feature.  
DC Regulators and Servos  
Closely related to DC and low frequency AC linear transducers are  
DC regulators and servo circuits. The proliferation of many battery,  
and solar powered remote instrumentation packages results in a  
need for adaptable circuits which may readily be made up from  
existing stock ICs. The examples given here are quite simple, but  
can be very useful to the designer when economy and size are at a  
premium.  
A variation on this particular idea is the remote strain gauge circuit  
operating from a three wire line, one of which is the shield. This  
full-differential input circuit has balanced  
input resistance to afford good common-mode noise rejection  
characteristics. Resistors are metal film or deposited carbon.  
Supply leads must be carefully bypassed close to the NE/SA5234  
with ceramic or chip monolithic capacitors to give optimum noise  
performance. As shown, an auxiliary sub-regulator may be added to  
improve the overall DC stability of the bridge signal voltage. A  
regulator capable of providing the necessary few milliamperes at  
somewhat reduced voltage for the transducer is shown in one of the  
following examples. This makes use of one of the op amps in the  
same device package to provide the voltage regulation. Note that  
the use of multiple op amps within a single package minimizes the  
possibility of thermal drift and mismatched response from various  
DC parameters.  
Solar Regulator for 3-Volt CMOS  
Working with small instrumentation packages which are to operate  
from solar photovoltaic cells may bring a need for simple  
sub-regulators for MOS circuits requiring only a few milliamperes of  
drain current. Figure 19 shows a simple low voltage regulator  
making use of the particularly excellent DC characteristics of the  
NE/SA5234. The regulator becomes an integral part of any  
functional analog signal processing package such as an  
environmental data instrumentation unit. The low current drain of  
the the typical 3V or 5V MOS digital IC allows one sub regulator to  
serve up to 10 or more such devices. If the instrument package is to  
be subjected to wide temperature variations, the SA5234 is  
recommended. A second op amp in the package may serve as a  
low battery alarm with tone modulator as in radio links, or simple  
logic level comparator. Overcurrent protection is easily added within  
the regulator loop to detect short circuit failures and automatically  
limit the current.  
Multiple sets of transducers may be constructed from The  
NE/SA5234 or the NE5234D surface mount device to form a  
compact and stable instrumentation package. This is useful for  
transducer applications in  
the measurement of pressure, strain, position and temperature,  
which have similar circuit configurations. First order temperature  
compensation of the transducers such as semiconductor strain  
gauges, or resistive units may be achieved by using one of the  
gauges as a reference device only. It is thermally coupled to the  
same member as the active gauge, as shown in the example.  
(Figure 18)  
DC Servo-amps  
Servo control systems for low voltage motor drives require high  
gain-accuracy and good DC stability for many applications.  
Applications such as the position control of air flow vanes, servo  
valves, and optical lenses or apertures, are typical examples.  
Figure 20 demonstrates one simple DC motor servo application with  
position control feedback. The motor is a 3V permanent magnet  
rotor type used in micro-position applications and is adaptable to  
battery supply environments.  
A 4 to 20mA Current Loop  
Some instrumentation installations require the 4-20mA current loop.  
This addition to the above bridge transducer circuit examples is  
demonstrated in Figure 16.  
Position information is received from a multi-turn potentiometer to  
give adequate resolution. The input voltage may be generated from  
another potentiometer which is remote from the motor drive unit  
proper, or from a D/A converter output for micro processor controlled  
systems. The input voltage range is 1.0 to 3.0V and the supply  
voltage is 4.5V.  
This circuit makes use of the remote transducer bridge previously  
described and adds current loop signaling capability. The  
voltage-to-current converter consists of an additional op amp from  
the same NE/SA5234 package combined with a single transistor to  
11  
 
1991 Oct  
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
20kΩ  
+
A2  
+4.5VDC  
10kΩ  
4
10Ω  
IC-1  
NE5234  
0.1µF  
VR1  
10Ω  
0.1µF  
BDX45  
V
CC  
+
+
+
150  
150  
A3  
A4  
A1  
IC-2  
NE5234  
100kΩ  
150  
150  
+
VR2  
+
VR2  
10kΩ  
A6  
A5  
R
S
C
L
BDX42  
V
REF  
C
PM  
MOTOR  
S
100  
1
VR1-3 = 1.4V  
SL00648  
Figure 20. Full Bridge Motor Drive  
audio impedance lines within a system. The use of two such  
amplifiers will provide stereo operation to +10dBm for a 600load.  
Active filters  
The NE5234 is easily adapted to use in a variety of active filter  
applications. Its high open-loop gain and excellent unity gain  
stability make it ideal for high-pass,  
Voice Operated Microphone  
The processing of voice transmissions for communications channels  
is generally coupled with the need for keeping the signal-to-noise  
ratio high and the intelligibility optimized for a given channel  
bandwidth. In addition, when a circuit is battery operated and  
portable, the requirement to obtain maximum battery life becomes  
important. The circuit example shown here is aimed at filling the  
need for a portable voice operated transmitter, cordless phone, or  
tape recorder. It utilizes the Philips Semiconductors NE5234 quad  
op amp in conjunction with the new low-voltage NE578 compandor  
to create an audio processor capable of operating in just such an  
environment. Both devices are operational to a low battery voltage  
of 2.0V. In addition the design further conserves current by  
automatically shifting the NE578 compandor to standby during the  
period when no transmissions are being made. Total current  
consumption at 3.0V is 2.8mA for the NE5234. In the active mode  
the NE578 draws 1.4mA and this drops to 170µA in the standby  
mode. This amounts to reducing the supply current demand by  
approximately 25% in the ‘listen mode’.  
band-pass and low-pass configurations operated with low voltage  
single supplies. Its low output impedance also makes it capable of  
obtaining low noise operation without resorting to separate high  
current buffers.  
Figure 21a shows the circuit for a VCVS low-pass filter with dual  
supply biasing and 600output termination. Figure 21b is a  
band-pass filter with AC coupled gain network for single supply  
operation.  
Communications and Audio  
Stereo Bridge Amplifier  
Figure 22 shows two NE5234 ICs in a bridge amplifier application.  
The choice of split supplies allows DC coupling, both from the input  
signal source and to the load. The gain is set to a nominal 20dB.  
Either inverting or non-inverting operation is available. The inverting  
input impedance is chosen as 600in order to match standard  
12  
 
1991 Oct  
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
Figure 23 shows the VOX audio circuit example. A description of  
its operation for voice activated transmission follows.  
DC control signal is fed to A4 which acts as a threshold comparator  
with extremely high gain and controlled hysteresis. This provides a  
positive going signal for releasing the NE578 from its inhibit mode  
when voice input is present. The NE578 is switched from standby  
mode when voice input is present. The NE578 is switched from  
standby mode to the active state by raising the voltage on Pin 8 of  
the device above 2V. Shutting the audio channel off requires this pin  
to be driven below 100mV. This demands the extremely wide output  
voltage swing of the NE5234 in order to reach this near to the  
negative rail voltage. The voltage threshold of the comparator, A4,  
Audio generated by the electret microphone is fed into the  
non-inverting input of preamp A1 and the signal amplified by 12dB.  
The biasing is accomplished by the resistive divider which provides  
a level of half the supply voltage which is connected through a 100k  
resistor to the non-inverting terminal of A1. This automatically  
provides ratiometric common mode biasing set at V /2 for the  
CC  
device. This level is then transferred directly to the following  
amplifier, A2, setting its DC operating point. The DC gain of both  
stage A1 and A2 are unity so the cumulative DC error is not  
multiplied by stage gain. The peak voice level is approximately  
is adjustable by use of the sensitivity control, R . It is used to allow  
S
the activation level to be raised or lowered depending upon the  
ambient audio level in the transmitter vicinity.  
100mV  
at the input to A1 from the microphone and this is  
RMS  
boosted to 400mV  
. The feedback network gain has a low  
RMS  
frequency corner at 160Hz and is flat up to the intersection of the  
closed loop gain with the open loop gain curve at nearly 500kHz.  
This would increase the noise bandwidth to an excessive degree  
unnecessary for voice channel communication. A band limiting  
network is, therefore, inserted across the feedback resistor to limit  
response to a nominal 5kHz.  
+3V  
–3V  
R
f
R
i
+
V
OUT  
V
IN  
R2  
R1  
600Ω  
Amplifier stage A2 is used to provide high level audio to the  
rectifier-filter stage for the rapid generation of a DC control signal for  
operating the voice activated switch function. Stage A2 gain is set  
to 20dB in order to allow activation of the voice channel on the rising  
edge of the first voice syllable. An attack time of 20ms is  
a. VCVS Low Pass Filter  
R
R
+5V  
implemented by adjusting the input charging impedance (R )  
S
between the rectifier and the A2 amplifier output. AC coupling must  
be used to isolate the DC common-mode voltage of the amplifier  
from the rectifier/storage capacitor and to allow only audio  
frequencies to drive the switching circuit. Amplifier A3 provides a  
high impedance unity gain buffer to allow a very slow decay rate to  
R5  
C1 C2  
R1  
C1  
+
V
IN  
V
OUT  
R3  
be applied to the time constant capacitor, C . The output of the  
R2  
R4  
T
C2  
storage capacitor reaches approximately 3.2V for a 250ms duration  
600Hz burst signal. Diode D1 (1N914) provides a negative clamp  
action which forces the full peak-to-peak voltage from A2 to charge  
the storage capacitor. D2 then acts to charge the capacitor to the  
peak input voltage minus one diode drop, 0.7V. Finally, the buffered  
b. VCVS Band Pass Filter  
SL00649  
Figure 21. Active Filters  
6kΩ  
10kΩ  
PIN  
+3V  
4
+
X(–1)  
600Ω  
10kΩ  
2
+
1
8
9
10  
AUDIO IN  
LEFT  
3
+
NE/SA5234  
#1  
11  
–3V  
13  
6
14  
7
5
12  
+
+
LEFT CHANNEL  
OUT  
BRIDGE AMP #2  
NE5234  
RIGHT CHANNEL  
OUT  
AUDIO IN  
RIGHT  
SL00650  
Figure 22. Stereo Bridge Amp  
13  
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
+4.5V  
10kΩ  
10kΩ  
10kΩ  
4
+
NE5234  
16  
19  
12  
100kΩ  
R
3
+
A2  
12kΩ  
+
A1  
1µF  
+4.5V  
R
A
2.2kΩ  
11  
10  
18kΩ  
MIC  
NE578  
SENS.  
ADJ.  
25kΩ  
0.15µF  
220Ω  
4.7µF  
R
S
D2  
0.47µF  
25kΩ  
D1  
C
1nF  
t
A4  
+
+
9
A3  
8
4.2V  
OFF  
7
R
D
10kΩ  
ON  
0V  
X1  
2.2MΩ  
11  
40.2kΩ  
SL00651  
Figure 23. VOX Audio System  
Other critical parameters in this type of circuit are the attack and  
decay times of the RC network which controls the operation of the  
voice operated switch. Attack time determines how quickly the  
circuit activates after a quiet period, and the decay time sets how  
long the transmitter channel stays active between words. It is  
important to reach an optimum balance between the two time  
constants in order to allow unbroken transmissions of good quality  
and no lost syllables. A 100 to 1 attack/decay ratio is used in this  
The compressor also has an attack time determined by capacitor C6  
on Pin 11. Attack time is 10k * C6, decay time equals four times this  
value. An auxiliary amplifier stage is used following the NE578 in  
order to allow bandwidth and special forms of equalization to be  
implemented. Note that 2:1 compression in a transmission will  
enhance the channel dynamic range and may be used with no  
further processing at the receiver, but feeding the received signal  
through the complimentary 2:1 expandor will achieve even greater  
enhancement of the recovered audio. The NE578 contains both  
operations in the same package. Please refer to Philips  
particular application and this is primarily set by the value of R and  
A
R . A typical delay of two seconds is easily accomplished. Due to  
D
extremely high input impedance of the buffer stage A3, R may be  
in the 1 to 2Mrange allowing a reasonable value of storage  
Semiconductors applications note AN1762 by Alvin K. Wong for  
complete information on these compandor circuits using the NE578.  
D
capacitor to be used.  
Fiber Optic Receiver for Low Frequency Data  
(Figure 26)  
The Audio Channel  
Audio input from the preamplifier, A1, is fed directly to Pin 14 of the  
NE578 compandor. Referring to Figure 24, which shows the  
internal diagram of the device, it can be seen that this is the  
compressor portion of the NE578. There is the option in this system  
to operate either in a 2:1 compressor mode or an automatic level  
control mode, (ALC). The compressor mode simply makes a 2:1  
reduction in the amplitude dynamic range of the input signal and  
brings it up to the chosen nominal 0dB output level which is  
This application makes use of the NE/SA5234 to detect photo-optic  
signals from either fiber or air transmitted IR (Infra-red) pulses. The  
signal is digitally encoded for the highest signal-to-noise ratio. The  
received signal is sensed by an IR photo diode which has its  
cathode biased to half the supply voltage (2.5V). The first gain  
stage is configured as a transimpedance amplifier to allow  
conversion from the microampere diode current signals to a voltage  
output of approximately 10mV . The second stage provides a  
0-P  
programmable from 10mV  
it is programmed for a 0dB level of 0.42V  
to 1V  
. In this particular example  
RMS  
RMS  
gain-of-ten amplifier to raise this signal level to 1V peak amplitude.  
This stage is directly coupled from the preamplifier stage in order to  
provide the necessary common-mode voltage of 2.5V. Its gain  
control network is capacitively coupled to prevent DC gain as is  
required in single supply configurations. Since this is essentially a  
pulse gain stage, low frequency gain below the signal repetition rate  
is not needed. The third stage acts in a limiting amplifier  
which is approximately  
RMS  
1V . This allows for a standardized output level with good  
P-P  
characteristics for FM modulation where peak deviation must be  
controlled. Figure 25 shows the input-output characteristics of the  
compressor and ALC.  
14  
 
1991 Oct  
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
configuration and its output is squared to swing approximately 5V,  
the standard TTL level. Again common-mode biasing is passed  
along from each of the stages up to the last in order minimize parts  
and simplify circuit layout. The final stage is a simple buffer  
amplifier to allow the receiver to drive a low impedance long wire  
line of 600to 900resistance. Some rise time response  
adjustment may be required. This is easily achieved following stage  
switching operation of the stage. However, care must be taken not  
allow the network’s time constant to become code dependent as to  
the average low frequency signal components or errors will result in  
the output signal.  
The advantage of this particular circuit is that it has the simplicity of  
single supply operation along with the capability of a large output  
swing making it fully TTL compatible  
three by using R -C to limit the rate of change of the signal voltage  
T
T
prior to the buffer. Note that the last stage acts as a zero-crossing  
detector. This maximizes noise immunity by allowing a transition  
REFERENCES:  
only after the third stage output voltage has risen above 2/3V  
.
CC  
Phase inversion may be accomplished, if the logic level signals are  
polarity reversed, by making stage 3 inverting and AC coupling the  
input signal with a sufficiently large capacitor to reduce droop.  
Stage 3 must then be biased by connecting its non-inverting node to  
bias point ‘A. This provides a 2.5V threshold for the proper  
Philips Semiconductors. Linear Data Manual, Volume 2 : Industrial.  
Sunnyvale: 1988.  
Wong, Alvin K. Companding with the NE577 and NE578..Philips  
Semiconductors Applications Note AN1762 : September 1990.  
10k  
GAINCELL  
C1  
EXP  
10k  
IN  
+
16  
V
1
2
3
4
5
6
7
8
G  
CC  
Σ
10µF  
R1*  
10k  
RECT.  
COMP  
CAP2  
5k  
15  
C8  
+
RECT  
2.2µF  
IN  
C7  
30k  
30k  
10k  
EXP  
+
+
CAP  
2.2µF  
14  
13  
12  
11  
10  
COMP  
IN  
C2  
10µF  
COMP  
EXPANDOR  
C3  
CAP1  
+
+
C6  
Σ
10µF  
EXP  
OUT  
C4  
2.2µF  
RECT.  
10k  
10k  
8.6k  
V
REF  
+
RECT  
IN  
10µF  
V
R3*  
REF  
BANDGAP  
V
GCELL  
G  
IN  
CC  
I
COMP.  
REF  
+
R2*  
PWRDN  
GND  
C5  
GAINCELL  
10µF  
MUTE  
+
COMP  
OUT  
C9  
10µF  
GND  
1nF  
C11  
TO  
PIN 4  
SUM  
+
PWRDN/  
MUTE  
9
IN  
NE578  
R4  
C10  
10µF  
*R1, R2 and R3 are 1% resistors.  
SL00652  
Figure 24. Block Diagram of NE578 Test and Application Circuit  
15  
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
[ ]12  
[ ]2  
REL LEVEL  
ABS LEVEL  
dBM  
INPUT TO G  
AND RECT  
DB  
V
COMPRESSION  
IN  
EXPANDOR  
OUT  
RMS  
(COMPRESSOR  
OUT)  
(EXPANDOR  
IN)  
A
D
B
C
+16dB  
0dB  
2.65V  
1.67V  
+16.0  
+10.68  
+12.0  
+6.68  
420mV  
42mV  
4.2mV  
420µV  
42µV  
0.0  
-5.32  
-20dB  
-40dB  
-60dB  
-80dB  
-20  
-40  
-60  
-80  
-25.32  
-45.32  
-65.32  
-85.32  
TRANSMISSION  
MEDIUM  
SL00653  
Figure 25. NE570/571/SA571 System Level  
+5V  
A
4
C
S
10mV  
R
+
1V  
+
100k  
I
10k  
O
R
t
1k  
R
1
2/3 V  
CC  
+V  
CC  
+5V  
T
+5V  
+
R
+
1.0  
C
T
5k  
1k  
SL00654  
Figure 26. Fiber Optic Data Receiver  
16  
1991 Oct  
 
Philips Semiconductors  
Application note  
Using the NE/SA5234 amplifier  
AN1651  
+3V  
4
1N9683  
8
6
7
+
9
+
5
10  
NE5234  
M
1N9683  
13  
12  
1
2
3
+
14  
+
11  
-3V  
1/100  
SL00655  
Figure 27. Half Bridge Servo  
17  
1991 Oct  
 

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